The present invention relates to a switching power supply circuit for use as a power supply for various electronic devices.
Switching power supply circuits that are widely known in the prior art employing a switching converter such as a flyback converter or a forward converter. These switching converters suffer a limitation on the suppression of switching noise because the switching waveform is rectangular. It is also known that their operating characteristics pose a limitation on efforts to increase the power conversion efficiency.
The applicant of the present application has proposed various switching power supply circuits that employ various resonant converters. The resonant converters are capable of easily achieving a high power conversion efficiency and are subject to low noise because the switching waveform becomes sinusoidal. The resonant converters are also advantageous in that they can be constructed of a relatively small number of parts.
FIG. 10 of the accompanying drawings shows an embodiment of a switching power supply circuit based on an invention proposed by the applicant.
The switching power supply circuit shown in FIG. 10 is capable of a maximum load power of 150 W or higher in a commercial power supply system of AC 100 V available in Japan and the U.S.A.
The switching power supply circuit shown in FIG. 10 has a voltage doubler rectifying circuit comprising rectifying diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 as a rectifying and smoothing circuit for rectifying and smoothing a commercial AC power supply AC. The voltage doubler rectifying circuit generates a DC input voltage 2Ei which is about twice a DC input voltage Ei that corresponds to the peak value of the AC power supply AC. If the AC input voltage VAC of the AC power supply AC is VAC=144 V, for example, then the DC input voltage 2Ei is about 400 V.
The voltage doubler rectifying circuit is used as the rectifying and smoothing circuit in order to meet the relatively high load condition of the maximum load power of 150 W or higher in the commercial power supply system of AC 100 V. Stated otherwise, since the DC input voltage is about twice the ordinary DC input voltage, the amount of a current flowing into a switching converter in a following stage is suppressed to keep the components of the switching power supply circuit reliable.
A rush current limiting resistor Ri is inserted in a rectifying current path of the voltage doubler rectifying circuit for suppressing a rush current which flows into smoothing capacitors when the power supply is turned on.
In FIG. 10, tne switching power supply circuit includes a self-excited voltage-resonant switching converter including a switching element Q1 which comprises a high-withstand-voltage bipolar transistor (BJT: junction transistor).
The switching element Q1 has a base connected to the positive terminal of the smoothing capacitor Ci1 (the rectified and smoothed voltage 2Ei) via a starting resistor RS, so that a base current will be produced from the rectifying and smoothing line when the switching power supply circuit starts to operate. A resonant circuit for self-excited oscillation, which comprises a series-connected circuit of an inductor LB, a detecting drive winding NB, a resonant capacitor CB, and a base current limiting resistor RB, is connected between the base of the switching element Q1 and primary-side ground.
A clamping diode DD inserted between the base of the switching element Q1 and the negative terminal of the smoothing capacitor C1 (primary-side ground) forms a path of a damper current which flows when the switching element Q1 is turned off. The switching element Q1 has a collector connected to a terminal of a primary winding N1 of a crossed insulated converter transformer PIT. The emitter of the switching element Q1 is connected to the ground.
A parallel resonant capacitor Cr is connected parallel between the collector and emitter of the switching element Q1. The capacitance of the parallel resonant capacitor Cr, and the combined inductance (L1+Lc) obtained by a series-connected circuit of the leakage inductance L1 of the primary winding N1 of the crossed insulated converter transformer PRT and the inductor Lc of a choke coil PCC jointly make up a primary-side parallel resonant circuit of the voltage-resonant converter. Although not described in detail here, when the switching element Q1 is turned off, the parallel resonant circuit causes a voltage Vcr across the resonant capacitor Cr to have a sine-wave pulse waveform, resulting in voltage-resonant operation.
The choke coil PCC has the inductor Lc and the detecting drive winding NB connected to each other as a transformer. The detecting drive winding NB induces an alternating voltage corresponding to a switching period due to a switching output transmitted from the primary winding N1 of the crossed insulated converter transformer PRT to the inductor Lc.
The crossed insulated converter transformer PRT has a function to transmit the switching output of the switching element Q1 to a secondary side thereof and also to perform constant-voltage control on the secondary-side output.
As shown in FIG. 11 of the accompanying drawings, for example, the crossed insulated converter transformer PRT comprises a three-dimensional core 200 having two double-C-shaped cores 201, 202 each with four magnetic legs, the magnetic legs of the double-C-shaped cores 201, 202 being connected at ends thereof to each other. The insulated converter transformer PRT has a primary winding N1 and a secondary winding N2 which are wound around two magnetic legs of the three-dimensional core 200 in one direction, and a control winding NC wound around two magnetic legs of the three-dimensional core 200 perpendicularly to the primary winding N1 and the secondary winding N2. The crossed insulated converter transformer PRT is thus constructed as a saturable reactor. The mating ends of the magnetic legs of the double-C-shaped cores 201, 202 are joined to each other with no gap defined therebetween.
The primary winding N1 of the crossed insulated converter transformer PRT has a terminal connected to the collector of the switching element Q1 and another terminal connected to the positive terminal of the smoothing capacitor C1 (the rectified and smoothed voltage 2Ei) via the series-connected inductor Lc of the choke coil PCC.
An alternating voltage is induced across the secondary winding N2 of the crossed insulated converter transformer PRT by the primary winding N1 thereof. A secondary-side parallel resonant capacitor C2 is connected parallel to the secondary winding N2, and the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary-side parallel resonant capacitor C2 jointly make up a parallel resonant circuit, which causes the alternating voltage to be induced as a resonant voltage across the secondary winding N2, thereby providing voltage resonant operation on the secondary side.
To the parallel resonant circuit on the secondary side, there are connected rectifying diodes D01, D02, D03, D04 and smoothing capacitors C01, C02 as shown via central taps of the secondary winding N2. The rectifying diodes D01, D02 and the smoothing capacitor C01 make up a full-wave rectification circuit, and the rectifying diodes D03, D04 and the smoothing capacitor C02 make up another full-wave rectification circuit.
The full-wave rectification circuit which is constructed of the rectifying diodes D01, D02 and the smoothing capacitor C01 is supplied with the resonant voltage from the parallel resonant circuit on the secondary side and generates a DC output voltage E01. The full-wave rectification circuit which is constructed of the rectifying diodes D03, D04 and the smoothing capacitor C02 is supplied with the resonant voltage from the parallel resonant circuit on the secondary side and generates a DC output voltage E02.
The DC output voltage E01 and the DC output voltage E02 are applied separately to a control circuit 1. The control circuit 1 employs the DC output voltage E01 as a detected voltage and the DC output voltage E02 as an operating power supply of the control circuit 1.
The control circuit 1 supplies a direct current whose level is variable depending on the level of the DC output voltage E01 to the control winding NC of the crossed insulated converter transformer PRT for thereby performing constant-voltage control as follows:
For example, when the AC input voltage VAC or the DC output voltage E01 varies due to a fluctuation of load power, the control circuit 1 changes a control current flowing through the control winding NC within a predetermined range.
Since the control winding NC is wound in the crossed insulated converter transformer PRT, leakage inductances L1, L2 can vary in the crossed insulated converter transformer PRT which is a saturable reactor.
As described above, the leakage inductance L1 of the primary winding N1 forms the primary-side parallel resonant circuit, and the leakage inductance L2 of the secondary winding N2 forms the secondary-side parallel resonant circuit. When the control current flowing through the control winding NC changes, the inductances L1, L2 are varied. Since the resonant impedances of the primary and secondary sides are also varied by the varying inductances L1, L2, the switching output transmitted from the primary side to the secondary side also varies for thereby making the secondary-side DC voltages E01, E02 constant. The above constant-voltage control process will hereinafter be referred to as a "parallel resonant frequency control process".
FIG. 12 of the accompanying drawings shows another switching power supply circuit based on an invention proposed by the applicant. The switching power supply circuit shown in FIG. 12 is also compatible with a maximum load power of 150 W or higher in a commercial power supply system of AC 100 V available in Japan and the U.S.A., as with the switching power supply circuit shown in FIG. 10. The switching power supply circuit includes a self-excited voltage-resonant switching converter including a switching element Q1 on a primary side.
Those parts in FIG. 12 which are identical to those shown in FIG. 10 are denoted by identical reference characters, and will not be described in detail below.
The switching power supply circuit shown in FIG. 12 has a crossed control transformer PRCT. As shown in FIG. 13, the crossed control transformer PRCT comprises a three-dimensional core 200 having two double-C-shaped cores 201, 202 each with four magnetic legs, the magnetic legs of the double-C-shaped cores 201, 202 being connected at ends thereof to each other. The crossed control transformer PRCT also comprises a controlled winding NR having a predetermined number of turns wound around two magnetic legs of the three-dimensional core 200 and a control winding NC wound around two magnetic legs of the three-dimensional core 200 perpendicularly to the controlled winding NR. The crossed control transformer PRCT is thus constructed as a saturable reactor.
The crossed control transformer PRCT can be regarded as a variable-inductance element, and is smaller in size than the crossed insulated converter transformer PRT shown in FIG. 11.
The controlled winding NR is inserted between the positive terminal of the smoothing capacitor Ci1 and the primary winding N1 of an insulated converter transformer PIT. In the power supply circuit shown in FIG. 12, a combined inductance (L1+LR) of a series-connected circuit of the leakage inductance L1 of the primary winding N1 side of the insulated converter transformer PIT and the inductance LR of the controlled winding NR, and the capacitance of the parallel resonant capacitor Cr jointly make up a parallel resonant circuit which performs primary-side switching operation as voltage-resonant operation.
As shown in FIG. 14, the insulated converter transformer PIT has an EE-shaped core 100 comprising two E-shaped cores 101, 102 of ferrite. No gap is defined between central magnetic legs of the E-shaped cores 101, 102. A primary winding N1 (and a detecting drive winding NB) and a secondary winding N2 are separately wound on the central magnetic legs, with separate bobbins actually used. The primary winding N1 and the secondary winding N2 provide a loose coupling of a coupling coefficient k.apprxeq.0.9.
A mutual inductance M of the insulated converter transformer PIT, due to the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2, may be +M (additive polarity mode) or -M (subtractive polarity mode) depending on the relationship between the polarities (winding directions) of the primary winding N1 and the secondary winding N2 and the connection of the rectifying diode D0 (D01, D02).
When the insulated converter transformer PIT operates in the connection shown in FIG. 15(a) of the accompanying drawings, the mutual inductance M becomes +M, and when the insulated converter transformer PIT operates in the connection shown in FIG. 15(b), the mutual inductance M becomes -M. In the illustrated circuits, the polarities of the primary winding N1 and the secondary winding N2 are in the additive polarity mode.
A secondary-side parallel resonant capacitor C2 is connected parallel to the secondary winding N2 of the insulated converter transformer PIT, providing a parallel resonant circuit.
The parallel resonant circuit causes the alternating voltage to be induced as a resonant voltage across the secondary winding N2. The resonant voltage is applied to two half-wave rectifying circuits, one comprising a rectifying diode D01 and a smoothing capacitor C02 and one comprising a rectifying diode D02 and the smoothing capacitor C02. The half-wave rectifying circuits produce respective DC output voltages E01, E02.
The rectifying diodes D01, D02 of the half-wave rectifying circuits are of the high-speed type because of rectifying an alternating voltage in a switching period.
For example, when the AC input voltage VAC or the DC output voltage E02 on the secondary side varies due to a fluctuation of load power, the control circuit 1 changes a control current flowing through the control winding NC within a range from 10 mA to 40 mA, for example. The inductance LR of the controlled winding NR thus changes in a range from 0.1 mH to 0.6 mH, for example.
Inasmuch as the controlled winding NR forms the parallel resonant circuit for providing voltage-resonant switching operation, the resonant condition of the parallel resonant circuit varies with respect to the switching frequency which is fixed. A sine-wave resonant pulse is produced across the parallel-connected circuit of the switching element Q1 and the parallel resonant capacitor Cr in an off period of the switching element Q1 because of operation of the parallel resonant circuit. The duration of the resonant pulse is variably controlled when the resonant condition of the parallel resonant circuit varies. That is, PWM (Pulse Width Modulation) control operation is achieved with respect to the resonant pulse. The PWM control of the duration of the resonant pulse is equivalent to the control of the off period of the switching element Q1. Stated otherwise, the on period of the switching element Q1 is variably controlled under the condition of the fixed switching frequency. When the on period of the switching element Q1 is thus variably controlled, the switching output transmitted from the primary winding N1 forming the parallel resonant circuit to the secondary side is varied, and the output level of the DC output voltages E01, E02 on the secondary side is also varied for thereby making the DC output voltages E01, E02 constant. The above constant-voltage control process will hereinafter be referred to as a "primary-side voltage resonant pulse duration control process".
FIG. 16 shows still another switching power supply circuit based on an invention proposed by the applicant. Those parts in FIG. 16 which are identical to those shown in FIGS. 10 and 12 are denoted by identical reference characters, and will not be described in detail below.
In the power supply circuit shown in FIG. 16, the crossed control transformer PRCT has a controlled winding on the secondary side.
The controlled winding of the crossed control transformer PRCT comprises two controlled windings NR, NR1. The controlled winding NR is connected in series between a terminal of the secondary winding N2 and the anode of the rectifying diode D01, and the controlled winding NR1 is connected in series between a tap output of the secondary winding N2 and the anode of the rectifying diode D02. In this connection, a secondary-side parallel resonant circuit includes inductances LR, LR1 of the controlled windings NR, NR1.
In the case where the controlled windings NR, NR1 of the crossed control transformer PRCT are on the secondary side, the inductances LR, LR1 of the controlled windings NR, NR1 are varied to variably control the pulse duration of a resonant voltage V2 across the secondary-side parallel resonant capacitor C2, i.e., the conduction angles of the secondary-side rectifying diodes. Constant-voltage control can be achieved by thus controlling the output level available on the secondary side.
The insulated converter transformer PIT provided to the power supply circuit has a structure which is the same as the structure shown in FIG. 14. The crossed control transformer PRCT has also a structure which is the same as the structure shown in FIG. 13 except that the controlled winding NR1 is additionally wound perpendicularly to the control winding NC. The crossed control transform r PRCT of the power supply circuit shown in FIG. 12 has the controlled winding NR inserted in the primary side and hence needs to have an insulation distance from the control winding NC that is connected to the secondary side in a DC manner, so that the crossed control transformer PRCT is required to have a corresponding size. However, the crossed control transformer PRCT of the power supply circuit shown in FIG. 16 may be smaller in size because the controlled winding NR1 is disposed in the secondary side and does not need to have an insulation distance from the control winding NC.
The switching power supply circuits shown in FIGS. 10 through 16 are compatible with the AC input voltage VAC of AC 100 V and the maximum load power of 150 W or higher, and hence the DC input voltage having a level of 2 Ei is produced by the voltage doubler rectifying circuit. Therefore, actually, a resonant voltage Vcr of 1800 V is generated across the switching element Q1 and the parallel resonant capacitor Cr when the switching element Q1 is turned off.
Therefore, the switching element Q1 and the parallel resonant capacitor Cr are required to be able to withstand the high voltage of 1800 V. Consequently, the switching element Q1 and the parallel resonant capacitor Cr are accordingly large in size. If a component of a high withstand voltage is used as the switching element Q1 in particular, then since a saturation voltage VCE(SAT), a storage time tSTG, and a fall time tf is large, and a current amplification factor hFE is small, it is difficult to set the switching frequency high. If the switching frequency is low, then the switching loss and the drive power are increased, resulting in a correspondingly increased power loss of the power supply circuit. The transformers in the power supply circuit and capacitors in the drive circuit are larger in size and more expensive, presenting an obstacle to efforts to reduce the size, weight, and cost of the power supply circuit.
In any of the constant-voltage control systems show in FIGS. 10, 12, and 16, the (crossed) insulated converter transformer PIT (PRT) which separates the primary side and the secondary side from each other has a certain coupling achieved in the absence of a gap, and the winding (inductor) Lc of the choke coil or the controlled winding NR of the crossed control transformer PRCT is connected in series to the primary winding N1 or the secondary winding N2. Therefore, the leakage inductance in the power supply circuit increases. The increase in the leakage inductance results in an increase in the leakage flux, possibly affecting an electric circuit as a load.
Actually, in order to reduce the effect of the leakage flux, the switching converter as a whole is housed in a shield case of aluminum having vent holes defined therein, and connected to input and output lines by connectors. Such a structure prevents the power supply circuit from being reduced in size, weight, and cost, and causes the power supply circuit to be manufactured for a considerably long period of time.